Double-Sided LCLC-Compensated Topology For Capacitive Power Transfer

ABSTRACT

A double-sided LCLC-compensated network is proposed for a capacitive power transfer (CPT) system. In one design, four metal plates are used to form two power transmitting and receiving capacitors and the LCLC network is used to compensate the capacitors. In the second design, two extra metal plates are used to couple with the previous four plates at the transmitting and receiving side, respectively, which forms the capacitor-integrated structure. The circuit parameter values are tuned to achieve zero voltage switching (ZVS) of the input side switches. There is also a CLLC topology proposed, which is a similar variation of LCLC circuit. A 3.3 kW input power capacitive power transfer prototype is designed and built. The experiment results show that the proposed CPT system can transfer 3.1 kW output power through an air gap distance of 70 mm with a dc-to-dc efficiency of 92.1%.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional Application No.62/141,437 filed on Apr. 1, 2015 and U.S. Provisional Application No.62/141,498 filed on Apr. 1, 2015. The entire disclosure of each of theabove applications is incorporated herein by reference.

GOVERNMENT CLAUSE

This invention was made with government support under Grant No.DE-EE0005565 awarded by the Department of Energy. The Government hascertain rights in this invention.

FIELD

The present disclosure relates to compensation circuits for capacitivepower transfer.

BACKGROUND

Capacitive power transfer (CPT) and inductive power transfer (IPT) aretwo effective methods to transfer power wirelessly. The CPT technologyutilizes high-frequency alternating electric fields to transfer powerwithout direct electric connection, while the IPT system uses magneticfield to transfer power. The IPT technology has already been widely usedin many applications, such as portable electronic devices, biomedicaldevices, and electric vehicle charging.

Compared with the IPT system, the CPT system has many advantages.Magnetic fields are sensitive to nearby metal objects and the systemefficiency drops quickly with this interference. They also generate eddycurrent losses and, hence, heat in a conductive object, which creates apotential fire hazard. However, the electric field in the CPT systemdoes not generate significant losses in the metal objects.

The recent CPT system can be classified by the matching networktopology. The most popular topology is a single inductor resonating withthe capacitor to form a simple series-resonant circuit. The secondtopology is the LCL structure at the front-end to step-up the voltagefor the coupling capacitor. However, there is also an inductor directlyconnected with the capacitor to form a series resonance. In these twotopologies, the series inductance is large because of the small value ofcapacitance. The voltage pressure on the capacitor is also large. Thethird topology is the resonant class E converter or the nonresonant PWMconverter, used to replace the compensation inductor. All of thesesystems require very high capacitance values, in the tens or hundreds ofnanofarad range. So, the transferred distance is usually around 1 mm.

This section provides background information related to the presentdisclosure which is not necessarily prior art.

DRAWINGS

The drawings described herein are for illustrative purposes only ofselected embodiments and not all possible implementations, and are notintended to limit the scope of the present disclosure.

FIG. 1 is a schematic depicting an example embodiment of a double-sidedLCLC compensation circuit in a wireless power transfer system;

FIG. 2A is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system;

FIG. 2B is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system excited only by the input source;

FIG. 2C is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system excited only by the output source;

FIGS. 3A and 3B are graphs illustrating input and output voltage as wellas the current waveform from a simulation of the wireless power transfersystem in FIG. 1;

FIG. 4 is a graph showing system DC-DC efficiency of the wireless powertransfer system at different misalignments;

FIG. 5A is a perspective view of the plates forming the couplingcapacitors;

FIGS. 5B and 5C are a front view and a side view, respectively, of theplates forming the coupling capacitors;

FIG. 5D is a schematic depicting the equivalent circuit of the platesforming the coupling capacitors;

FIG. 6A is a perspective view depicting an alternative integratedstructure of plates forming the coupled capacitors;

FIGS. 6B and 6C are a front view and a side view, respectively, of theintegrated structure of the plates;

FIG. 6D is a schematic depicting the equivalent circuit of theintegrated structure of plates;

FIG. 7 is a schematic depicting an example embodiment of a double-sidedCLLC compensation circuit in a wireless power transfer system;

FIG. 8 is a schematic of a circuit model of the coupling plates in thewireless power system of FIG. 7;

FIG. 9 is a schematic of the wireless power system of FIG. 7 with asimplified capacitor model;

FIG. 10A is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system of FIG. 7;

FIG. 10B is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system excited only by the input source;

FIG. 10C is a schematic illustrating a fundamental harmonic analysis ofthe wireless power transfer system excited only by the output source;

FIG. 11A is a perspective view of the plates forming the couplingcapacitors in the wireless power transfer system of FIG. 7;

FIG. 11B is a front view of the plates forming the coupling capacitorsin the wireless power transfer system of FIG. 7;

FIGS. 12A and 12B are graphs depicting capacitance value at differentmisalignment conditions;

FIGS. 13A and 13B are graphs depicting capacitance value at differentair gap conditions;

FIG. 14 is a diagram of a capacitive-coupled roadway power electricvehicle system;

FIG. 15 is a diagram of bottom of an electric vehicle equipped for usein the roadway power electric vehicle system;

FIGS. 16A and 16B are cross-sectional views of the capacitive-coupledroadway power electric vehicle system with the vehicle aligned andmisaligned, respectively; and

FIG. 17 is a block diagram of an example embodiment of thecapacitive-coupled roadway power electric vehicle system.

Corresponding reference numerals indicate corresponding parts throughoutthe several views of the drawings.

DETAILED DESCRIPTION

Example embodiments will now be described more fully with reference tothe accompanying drawings.

FIG. 1 depicts an example embodiment of a double-sided LCLC compensationcircuit 8 in a wireless power transfer system 10. The wireless powertransfer system is comprised generally of a send unit 12, a receive unit16 and a pair of coupling capacitors 15. In operation, the send unit 12is configured to transfer power capacitively through the pair ofcoupling capacitors 15 to the receive unit 16. In one example, thereceive unit 16 is integrated into a vehicle to support wirelesscharging of batteries therein. Other applications for the wireless powertransfer system also fall within the broader aspects of this disclosure.

The send unit 12 includes an inverter 13 and a send side compensationcircuit 14. The inverter 13 is configured to receive a DC input signaland converts the DC input signal to an AC input signal at a desiredresonant frequency. In the example embodiment, the inverter is a fullbridge converter circuit comprised of four switches. In anotherembodiment, the inverter may be a half-bridge converter circuitcomprised of two switches. Other types of inverter circuits also fallwithin the scope of this disclosure.

The send side compensation circuit 14 interconnects the inverter 13 withthe pair of coupling capacitors 15. Since the challenge in CPT system isbrought on by the small capacitance value, one way to solve it is toconnect extra capacitors in parallel with the coupling capacitor toincrease the capacitance in the resonant circuit. It follows that thesend side compensation circuit 14 includes at least two bypasscapacitors C₁, C_(f1), where each bypass capacitor is connected inparallel between input terminals of the pair of coupling capacitors.

In the example embodiment, the send side compensation circuit 14 iscomprised of two LC circuits coupled in series. In one LC circuit, afirst bypass capacitor C₁ is electrically coupled in parallel betweeninput terminals of the first and second coupling capacitors C_(s1),C_(s2), and the output terminal of a first inductor L₁ is electricallycoupled at a first node to an input terminal of the first couplingcapacitor C_(s1). In the other LC circuit, a second bypass capacitorC_(f1) is also electrically coupled in parallel with the first bypasscapacitor, and an output terminal of the second inductor L_(f2) iselectrically coupled at a second node to the input terminal of the firstinductor.

The receive 16 unit includes a receive side converter 18 and a receiveside compensation circuit 17. The converter 18 is configured to receivean AC charging signal from the pair of coupling capacitors 15 andconverts the AC charging signal to a DC charging signal. In an exampleembodiment, the converter 18 is a full wave rectifier circuit althoughother types of converter circuits are also contemplated by thisdisclosure. A battery or another type of load 19 may be configured toreceive the DC charging signal from the converter.

The receive side compensation circuit 17 interconnects the pair ofcoupling capacitors 15 with the receive side converter 18. Likewise, thereceive side compensation circuit 17 includes at least two bypasscapacitors C₂, C_(f2), where each bypass capacitor is connected inparallel between output terminals of the pair of coupling capacitors.

In the example embodiment, the receive side compensation circuit 17 issymmetric with the send side compensation circuit 14. That is, thereceive side compensation circuit 17 is comprised of two LC circuitscoupled in series. In one LC circuit, a third bypass capacitor C₂ iselectrically coupled in parallel between output terminals of the firstand second coupling capacitors C_(s1), C_(s2); and the input terminal ofthe third inductor L₂ is electrically coupled at a third node to theoutput terminal of the first coupling capacitor C_(s1). In the other LCcircuit, a fourth bypass capacitor C_(f2) is electrically coupled inparallel with the third bypass capacitor C₂; and the input terminal ofthe fourth inductor L_(f2) is electrically coupled at a fourth node tothe output terminal of the third inductor L₂.

First and third bypass capacitors C1 and C2 are connected in parallelwith the coupling capacitors C_(s1), C_(s2). As long as capacitance ofthe first and third bypass capacitors C1 and C2 are much larger than thecoupling capacitors, most of the current is bypassed by C1 and C2 toreduce the voltage stress. At the input side, the LCL topology is usedto step up the voltage for the capacitor. At the output side, the otherLCL topology steps down the voltage to the load.

In the example embodiment, the H-bridge inverter provides square waveexcitation voltage to the resonant circuit and the output side rectifierconverts the AC current to DC signal to supply the load. Since the LCLCnetwork works as a high frequency filter at both the input and output,the fundamental harmonic analysis (FHA) of the CPT system is sufficientto calculate the performance. The simplified circuit model is shown inFIG. 2A.

Using the superposition theory, the resonant circuit can be divided intotwo parts as shown in FIG. 2B and FIG. 2C. FIG. 2B shows the resonancebehavior in the circuit when only the input voltage V₁ is effective,which is defined as Mode I. FIG. 2C shows the resonance behavior in thecircuit when only the output voltage V₂ is effective, which is definedas Mode II. Since FIG. 2A is a linear circuit, the system performancecan be calculated by the add-up of FIG. 2B and FIG. 2C. Based on FIG. 2Band FIG. 2C, the parameter values satisfy the following equation (1).

$\begin{matrix}\left\{ \begin{matrix}{L_{f\; 1} = {1\text{/}\left( {\omega_{0}^{2}C_{f\; 1}} \right)}} \\{L_{f\; 2} = {1\text{/}\left( {\omega_{0}^{2}C_{f\; 2}} \right)}} \\{L_{1} = {{1\text{/}\left( {\omega_{0}^{2}C_{p\; 1}} \right)} + L_{f\; 1}}} \\{L_{2} = {{1\text{/}\left( {\omega_{0}^{2}C_{p\; 2}} \right)} + L_{f\; 2}}} \\{C_{p\; 1} = {C_{1} + {{C_{s} \cdot C_{2}}\text{/}\left( {C_{s} + C_{2}} \right)}}} \\{C_{p\; 2} = {C_{2} + {{C_{s} \cdot C_{1}}\text{/}\left( {C_{s} + C_{1}} \right)}}} \\{C_{s} = {{C_{s\; 1} \cdot C_{s\; 2}}\text{/}\left( {C_{s\; 1} + C_{s\; 2}} \right)}} \\{\omega_{0} = {2\; {\pi \cdot f_{sw}}}}\end{matrix} \right. & (1)\end{matrix}$

where, f_(sw) is the switching frequency and C_(s) is the series of thetwo coupling capacitors.

FIG. 2B shows that the output current only depends on the input voltageand the input current only depends on the output voltage. Moreover,since the H-bridge diodes rectifier is used at the secondary side, thecurrent and voltage are in phase. FIG. 2B indicates that the output sidecurrent is 90 degree leading the input voltage, and FIG. 2C indicatesthat the input side current is 90 degree lagging the output voltage.Therefore, the current and voltage are also in phase with each other atthe input side, which means the input side power factor is equal to 1.As a result, if the power losses in the components can be neglected, theinput power can be expressed as in equation (2) as follows

$\begin{matrix}{P_{in} = {{V_{1} \cdot I_{1}} = {\omega_{0}{C_{s} \cdot \frac{C_{f\; 1}C_{f\; 2}}{{C_{1}C_{2}} + {C_{f\; 1}C_{s}} + {C_{f\; 2}C_{s}}} \cdot V_{1} \cdot V_{2}}}}} & (2)\end{matrix}$

As mentioned above, C₁ and C₂ are used to bypass the current flowingthrough Cs. Their values should be chosen to be at least five (5) timesand preferably ten (10) times capacitance of C. As a result, the outputpower can be estimated as in equation (3).

$\begin{matrix}{P_{in} = {{V_{1} \cdot I_{1}} \approx {\omega_{0}{C_{s} \cdot \frac{C_{f\; 1}C_{f\; 2}}{C_{1}C_{2}} \cdot V_{1} \cdot V_{2}}}}} & (3)\end{matrix}$

Equation (3) shows that the input power is proportional to the couplingcapacitance, input voltage, and output voltage, which makes it mucheasier in the system parameter design.

For illustration purposes, a 3.3 kW input power CPT charging system isdesigned. The coupling capacitance can be calculated as in equation (4).

$\begin{matrix}{C_{s\; 1} = {C_{s\; 2} = {{ɛ_{0}ɛ_{r}\frac{A}{}} = {{8.85 \times {10^{- 12} \cdot \frac{1}{0.15}}} = {59\; {pF}}}}}} & (4)\end{matrix}$

where, ε₀ is the permittivity of vacuum, ε_(r) is the relativepermittivity of air, A is the effective coupling area for each capacitor(e.g., estimated to be 1 m²), and d is ground clearance of the vehicle(e.g., estimated to be 0.15 m). So the equivalent capacitance of thecoupling capacitor, which is the series of C_(s1) and C_(s2), iscalculated to be 29.5 pF.

In this example, the input and output dc voltage V_(in) and V_(out) areset to be 400V and the corresponding AC voltage at the input and outputside are V₁=V₂=2√{square root over (2)}/π×400=360V. The systemparameters are designed to be symmetric and all the parameter values aredesigned according to equation (1) and (2). The parameter values areshown in Table I.

TABLE I A 3.3 kW CPT System Parameter with 1 m²/0.15 m plates V_(in)V_(out) f_(sw) L_(f1) C_(f1) L₁ C₁ C_(s) C₂ L₂ C_(f2) L_(f2) 400 V 400 V1 MHz 11.86 μH 2.14 nF 156.9 μH 150 pF 29.5 pF 150 pF 164.7 μH 2.14 nF11.86 μHIn table I, the inductor L₂ is designed to be 5% larger than L₁ toprovide soft-switching condition to the input side H-bridge inverter.LTspice can be used to simulate the designed CPT system performance asshown in FIGS. 3A and 3B.

FIG. 3A shows that the input current I₁ is lagging the input voltage,which can maintain the soft-switching of the MOSFETs. In FIG. 3B, theoutput voltage is leading the input voltage, which agrees with theanalysis above.

In an example embodiment, since it is difficult to make four 1 m² platesto form the two capacitors, four 0.61 m×0.61 m aluminum plates areutilized and the distance is set to be 70 mm. In this embodiment, thecorresponding coupling capacitor of two plates is calculated to be 47 pFand the equivalent capacitance is 23.5 pF, which is not far from thedesired value of 29.5 pF. Consequently, parameters for the compensationcircuit parameter are set forth in Table II.

TABLE II A 3.3 kW CPT System Prototype Parameter with 0.37 m²/0.07 mplates V_(in) V_(out) f_(sw) L_(f1) C_(f1) L₁ C₁ C_(s) C₂ L₂ C_(f2)L_(f2) 400 V 400 V 1 MHz 10.4 μH 2.45 nF 145 μH 160 pF 23.5 pF 160 pF166 μH 2.45 nF 10.4 μH

FIG. 4 depicts system efficiency at different misalignments. In FIG. 4,the output power denotes the power received by the dc side load and theefficiency is from the input side dc source to the output side dc load.It is shown that there is very small power drop at 10 cm and 20 cmmisalignment conditions. Even when the misalignment increases to 30 cm,which is about half of the plate size, the output power remains 70% ofthe no-misalignment condition. This is much better than the previousinductive power transfer (IPT) system. In a IPT system with coil size600 mm×800 mm, the output power drops to about 50% at 310 mmmisalignment. FIG. 4 also shows another benefit that the outputefficiency changes a little at 10 cm and 20 cm misalignment and it dropsonly 1% at 30 cm misalignment case.

In one embodiment, the first and third bypass capacitors C₁ and C₂ aredesigned as separated capacitors and they are not integrated with thecoupling capacitors C_(s1), C_(s2). Thus, the coupling capacitorsC_(s1), C_(s2) are comprised of four metal plates as shown in FIG. 5A.Example dimensions for the coupling capacitors are given in FIG. 5A-5C.It is understood that dimensions for the capacitors will vary dependingon distance between plates, desired power output and other factors. Anequivalent capacitor model is shown in FIG. 5D. In this model, thecapacitor between P1 and P3 and the capacitor between P2 and P4 are thetwo coupling capacitors. When the distance d₂ between P1 and P2 is largeenough, the other coupling between the plates can be neglected.

In an alternative embodiment, the first and third capacitors C₁ and C₂are integrated with the coupling capacitors C_(s1), C_(s2) as shown inFIG. 6A. Example dimensions for the coupling capacitors is again givenin FIG. 6A-6C and an equivalent capacitor model is shown in FIG. 6D. Inthis integrated model, the capacitor between P1 and P3 and the capacitorbetween P2 and P4 are the two coupling capacitors. P5 is added at theprimary side to increase the capacitance between P1 and P2. The couplingcapacitor C₁₁ between P5 and P1 is in series with the capacitor C₁₂between P5 and P2, which forms the compensation capacitor C₁ in FIG. 1.P6 is added at the secondary side to increase the capacitance between P3and P4. The coupling capacitor C₂₁ between P6 and P3 is in series withthe capacitor C₂₂ between P6 and P4, which forms the compensationcapacitor C₂ in FIG. 1. Similarly, when the distance d₂ between P1 andP2 is large enough, the other coupling between the plates can beneglected. In this way, the external compensation capacitors C1 and C2can be eliminated;

The advantage is that the system structure is more compact and lessconnectors are needed in the installation process.

FIG. 7 depicts another embodiment of a compensation circuit 71, 72 in awireless power transfer system 10. As described above, the wirelesspower transfer system 10 is comprised generally of a send unit 12, areceive unit 16 and a pair of coupling capacitors 15. Two pairs of metalplates are used to form the two coupling capacitors in a loop totransfer power through a capacitive coupling. The compensation circuitresonates with the coupling capacitors to generate high voltage on theplates in order to achieve power transfer. On the primary side, a fullbridge inverter 13 is used to provide AC excitation to the resonant tankalthough other types of inverters may be used as well. On the secondaryside, a full-bridge diode rectifier 18 is utilized to provide dc voltageto the load although other types of rectifiers may be used as well.

In this embodiment, an CLLC topology is proposed for the compensationcircuit on both the primary side and the secondary side. For the sendside compensation circuit 71, a first bypass capacitor C_(ext1) iselectrically coupled in parallel between input terminals of the firstand second coupling capacitors C_(s1), C_(s2), and the output terminalof a first inductor L₁ is electrically coupled at a first node 74 to aninput terminal of the first coupling capacitor C_(s1). Additionally, asecond inductor L_(f1) is electrically coupled in parallel with thefirst bypass capacitor C_(ext1), and an output terminal of the secondcapacitor C_(f1) is electrically coupled at a second node 75 to theinput terminal of the first inductor L₁.

In this example embodiment, the receive side compensation circuit 72 issymmetric with the send side compensation circuit 71. That is, a CLLCtopology also proposed for the receive side compensation circuit 72.Specifically, a third bypass capacitor C_(ext2) is electrically coupledin parallel between output terminals of the first and second couplingcapacitors C_(s1), C_(s2); and the input terminal of the third inductorL₂ is electrically coupled at a third node 76 to the output terminal ofthe first coupling capacitor C_(s1). A fourth inductor L_(f2) iselectrically coupled in parallel with the third bypass capacitorC_(ext2); and the input terminal of the fourth capacitor C_(f2) iselectrically coupled at a fourth node 77 to the output terminal of thethird inductor L₂.

The circuit model of the coupling plates should be derived to design thecompensation circuit parameters. In one embodiment, the two pairs ofplates are arranged 500 mm away from each other, and the cross-couplingbetween the two pairs is neglected. In this design, the couplers areplaced closer together to make the system more compact. Therefore, theinter-coupling should be considered and modeled. The six couplingcapacitors between the plates and the equivalent circuit model with fourcapacitors are shown in FIG. 8. The simplification process will beprovided. If a current source IP1 is applied at the plates P1 and P2,the nodal current equation is expressed as,

$\begin{matrix}\left\{ \begin{matrix}\begin{matrix}{{{\left( {C_{12} + C_{13} + C_{14}} \right) \cdot V_{P\; 1}} - {C_{13} \cdot V_{P\; 3}} - {C_{14} \cdot V_{P\; 4}}} = {I_{P\; 1}\text{/}\left( {j\; \omega_{0}} \right)}} \\{{{{- C_{13}} \cdot V_{P\; 1}} + {\left( {C_{13} + C_{23} + C_{34}} \right) \cdot V_{P\; 3}} - {C_{34} \cdot V_{P\; 4}}} = 0}\end{matrix} \\{{{{- C_{14}} \cdot V_{P\; 1}} - {C_{34} \cdot V_{P\; 3}} + {\left( {C_{14} + C_{24} + C_{34}} \right) \cdot V_{P\; 4}}} = 0}\end{matrix} \right. & (5)\end{matrix}$

where V_(P1), V_(P2), V_(P3), and V_(P4) are the voltage on each plate(V_(P2)=0 is set to be the reference node), ω₀=2πf_(sw), f_(sw) is theswitching frequency, and I_(P1) is the fundamental external inputcurrent flowing into P₁. In order to simplify the circuit model, therelationship between the plate voltage is derived.

$\begin{matrix}\left\{ \begin{matrix}{V_{P\; 3} = {\frac{{C_{13}C_{14}} + {C_{13}C_{24}} + {C_{13}C_{34}} + {C_{14}C_{34}}}{{C_{34}\left( {C_{13} + C_{14} + C_{23} + C_{24}} \right)} + {\left( {C_{13} + C_{23}} \right)\left( {C_{14} + C_{24}} \right)}} \cdot V_{P\; 1}}} \\{V_{P\; 4} = {\frac{{C_{13}C_{14}} + {C_{14}C_{23}} + {C_{13}C_{34}} + {C_{14}C_{34}}}{{C_{34}\left( {C_{13} + C_{14} + C_{23} + C_{24}} \right)} + {\left( {C_{13} + C_{23}} \right)\left( {C_{14} + C_{24}} \right)}} \cdot V_{P\; 1}}}\end{matrix} \right. & (6)\end{matrix}$

Considering (6) and the first equation in (5), the equivalent inputcapacitance, C_(in)=I₁/(jω₀V_(P1)), seen from the P₁ and P₂ side isexpressed as,

$\begin{matrix}{C_{in} = {C_{12} + \frac{{C_{34}\left( {C_{13} + C_{14}} \right)}\left( {C_{23} + {C\; 24}} \right)}{{C_{34}\left( {C_{13} + C_{14} + C_{23} + C_{24}} \right)} + {\left( {C_{13} + C_{23}} \right)\left( {C_{14} + C_{24}} \right)}} + \frac{{C_{13}{C_{23}\left( {C_{14} + C_{24}} \right)}} + {C_{14}{C_{24}\left( {C_{13} + C_{23}} \right)}}}{{C_{34}\left( {C_{13} + C_{14} + C_{23} + C_{24}} \right)} + {\left( {C_{13} + C_{23}} \right)\left( {C_{14} + C_{24}} \right)}}}} & (7)\end{matrix}$

For the plates, the voltage between P1 and P2 is treated as the input,and the voltage between P3 and P4 is treated as the output. The transferfunction between the two voltages can be defined as H=(VP3−VP4)/VP1.Considering (6), the transfer function H is expressed as,

$\begin{matrix}{H = \frac{{C_{13}C_{24}} - {C_{14}C_{23}}}{{C_{34}\left( {C_{13} + C_{14} + C_{23} + C_{24}} \right)} + {\left( {C_{13} + C_{23}} \right)\left( {C_{14} + C_{24}} \right)}}} & (8)\end{matrix}$

The plates structure can be designed to be symmetric between the primaryand secondary sides. For the equivalent model in FIG. 8, there existsC_(s1)=C_(s2), and C_(int1)=C_(int2). The input capacitance and transferfunction are expressed as,

$\begin{matrix}\left\{ \begin{matrix}{C_{in} = {C_{{int}\; 1} + \frac{C_{s} \cdot C_{{int}\; 2}}{C_{s} + C_{{int}\; 2}}}} \\{H = \frac{Cs}{{Cs} + C_{{int}\; 2}}} \\{{Cs} = \frac{C_{s\; 1} \cdot C_{s\; 2}}{C_{s\; 1} + C_{s\; 2}}}\end{matrix} \right. & (9)\end{matrix}$

The equivalent capacitors can be expressed as,

$\begin{matrix}\left\{ \begin{matrix}{C_{{int}\; 1} = {C_{in} \cdot \frac{1}{1 + H}}} \\{{Cs} = {C_{in} \cdot \frac{H}{1 - H^{2}}}}\end{matrix} \right. & (10)\end{matrix}$

Therefore, using (8), (9), and (10), the equivalent capacitor model canbe derived for any given plates' dimensions.

Replace the four plates in FIG. 7 with the equivalent circuit model inFIG. 8. The circuit topology is re-drawn in FIG. 9. The CLLC topology isused to work with the plates. The external capacitors C_(ext1) andC_(ext2) are connected in parallel with the internal capacitors C_(int1)and C_(int2). The total compensation capacitance can be defined asC=C₁=C₂=C_(ext1)+C_(int1)=C_(ext2)+C_(int2). The compensation parametersare also designed to be symmetric, then L_(f)=L_(f1)=L_(f2),C_(f)=C_(f1)=C_(f2). The fundamental harmonics approximation (FHA) canbe used to analyze the working principle of the circuit at the resonantfrequency. The input and output square wave sources are treated assinusoidal source as shown in FIG. 10A. The superposition theorem isused to analyze the two sources separately.

FIG. 10B shows the circuit is excited only by the input source. The tworesonances can be expressed as,

$\begin{matrix}\left\{ \begin{matrix}{L_{f\; 2} = \frac{1}{\omega_{0}^{2} \cdot C_{f\; 2}}} \\{L_{1} = {\frac{1}{\omega_{0}^{2} \cdot C_{in}} - L_{f\; 1}}}\end{matrix} \right. & (11)\end{matrix}$

The output current on C_(f2) can be expressed as,

$\begin{matrix}{I_{2} = {\frac{L_{f\; 1} + L_{1}}{L_{f\; 1}} \cdot \frac{C_{s}}{C_{s} + C_{2}} \cdot \frac{V_{1}}{j\; \omega_{0}L_{f\; 2}}}} & (12)\end{matrix}$

Similarly, FIG. 10C shows the circuit is excited only by the outputsource. The two resonances can be expressed as,

$\begin{matrix}\left\{ \begin{matrix}{L_{f\; 1} = \frac{1}{\omega_{0}^{2} \cdot C_{f\; 1}}} \\{L_{2} = {\frac{1}{\omega_{0}^{2} \cdot C_{in}} - L_{f\; 2}}}\end{matrix} \right. & (13)\end{matrix}$

Also, the input current on C_(f1) can be expressed as,

$\begin{matrix}{I_{1} = {\frac{L_{f\; 2} + L_{2}}{L_{f\; 2}} \cdot \frac{C_{s}}{C_{s} + C_{1}} \cdot \frac{V_{2}}{j\; \omega_{0}L_{f\; 1}}}} & (14)\end{matrix}$

Since a full-bridge diode rectifier is used on the secondary side, theoutput voltage and current are in phase with each other. Consideringequations (11)-(14), the output power can be expressed as,

$\begin{matrix}{P_{out} = {{{V_{2}} \cdot {{- I_{2}}}} = {\frac{{\omega_{0} \cdot C_{s} \cdot C_{f\; 1}}C_{f\; 2}}{{C_{1}C_{2}} + {C_{1}C_{s}} + {C_{2}C_{s}}} \cdot {V_{1}} \cdot {V_{2}}}}} & (15)\end{matrix}$

As compared to the double-sided LCLC topology described above, theoutput power of the CLLC compensated system is the same as that of theLCLC system. The inductances of L₁ and L₂ can be reduced to make iteasier to implement them.

Dimensions for an example embodiment of the capacitive coupler are shownin FIGS. 11A and 11B. The area of the plate determines the couplingcapacitance. Although different shapes are contemplated, each plate hasa square shape, while the length l₁ is 24 in (610 mm). The two pairs areseparated, and the distance between them, d_(c), is 300 mm. The air gap,d, is 150 mm. The thickness of the plates does not relate to thecoupling capacitance and it is set to be 2 mm.

Finite element analysis (FEA) by Maxwell is used to determine thecapacitance matrix that contains the coupling capacitance between eachpair of plates. Based on the FEA results, the equivalent capacitances,C_(int1), C_(int2), C_(s1), and C_(s2) can be calculated using equations(7)-(10). The misalignment ability is also an important designspecification. The X, Y, and Z directions are indicated in FIG. 11A.When there is misalignment in either X or Y direction, the variation ofthe coupling capacitance is as shown in FIGS. 12A and 12B. It shows thatC_(s) decreases with the increasing misalignment. Since equation (15)shows that the system output power is proportional to C_(s), it meansthe system power will decrease with misalignment. For the othercapacitor, C_(int1), which is used to resonate with L₁, its valueincreases with misalignment. However, its variation is relatively smalland its influence on the system power can be neglected. When comparingthe X and Y direction misalignments, FIGS. 12A and 12B also show thatthe capacitances are more sensitive to Y direction misalignment.

The variation of air gap distance is also studied, and the capacitancesare shown in FIGS. 13A and 13B. FIG. 13A shows that when the air gapdistance increases from 150 mm to 300 mm, the coupling capacitance,C_(s) reduces by one half, which means that the system power will alsoreduce by half. The other capacitance, C_(int1) is not sensitive to theair gap variation, hence it only changes by about 15%.

After the coupler structure and compensation circuit topology have beendesigned in the previous sections, a 2.9 kW input power CLLC-compensatedCPT system is designed according to the power requirement in equation(15). The parameter values are calculated using equations (11) and (13).All the system specifications and circuit parameter values are shown inTable 3 below.

TABLE 3 System Specifications and Circuit Parameters Parameter DesignValue Parameter Design Value V_(in) 400 V V_(out) 450 V I₁ 610 mm d 150mm f_(sw) 1 MHz C_(s) 14.0 pF L_(f1) 11.76 μH L_(f2) 11.76 μH C_(f1)2.15 nF C_(f2) 2.15 nF L₁ 164.0 μH L₂ 165.8 μH C₁ 130 pF C₂ 130 pF

The input dc voltage is 400 V, and the output dc voltage is 450V torepresent the battery pack on the vehicle side. Since the system poweris proportional to the switching frequency, the frequency is set to be 1MHz to increase the output power. Compared to double-sided LCLC topologydescribed above, compensation inductor L₁ is decreased from 231 pH to164 pH, which is easier to make in practice. It also needs to beemphasized that inductor L₂ is designed to be larger than L₁ to providesoft-switching condition to the input side inverter.

In another aspect of this disclosure, a capacitive-coupled roadway powerelectric vehicle system 100 is presented. The capacitive-coupled roadwaypower electric vehicle system 100 includes: (1) an electric vehicle(EV); and (2) a roadway network over which the vehicle travels. Theelectric vehicle includes onboard energy storage devices that canrapidly recharged or energized with energy obtained from an electricalcurrent. The electric vehicle further includes an energy receivingdevice. The energy storage device of the vehicle will be charged whilethe vehicle is in operation. The roadway network includes a network ofroadway electric power sending modules that have been electrified with amultiplicity of roadway power segments embedded in or on the roadway.The EV can be recharged while the EV is moving on the roadway. As thevehicle passes over such capacitive-coupling power sending roadway,electric power is coupled to the electric vehicle through the sendingplates in the roadway to the receiving plates mounted on the chassis ofthe vehicle through the electric field.

Referring to FIG. 14, an electric vehicle (EV) 130 is shown traversingin a capacitive-coupled roadway power system 100 with power sendingplates thereon made in accordance with the present disclosure. Asillustrated in FIG. 1, the electric vehicle 130 is of a conventionalform having a conventional wheel system for support of the vehicle 130above the road surface. In one embodiment, the roadway energy sendingpart includes a pair of road surface mounted metal plates with highconductivity 110 a, 110 b. The two sending plates are distributedbetween the left edge of the road 140 to the right edge of the road 150.There should be some distance between one pair of sending plates. Thesending plates are separated into segments 110, 120 that are preferablydisposed parallel to the direction of travel of a vehicle, such asvehicle 130, on the powered roadway system.

In one embodiment, the roadway power EV system 100 has one power source160 for each power sending segment. Each power sending segment 110, 120can power one or more EVs 130. The power requirement of each segment isdecided by the number of vehicles 130 capacitively coupled to thesegment. In other embodiments, a power source may power multiple roadsegments.

A bottom view of the receiving plates mounted on the chassis of anelectric vehicle is illustrated in FIG. 15. In one embodiment, theenergy receiving part on the vehicle chassis includes a pair of plates210 a, 210 b with the material of high conductivity that are distributedin either side of a central axis of the EV chassis. There should be somedistance between the pair of receiving plates 210 a, 210 b. Since themajority of EV chassis contains metal material, a layer of isolationmaterial 220 with high resistivity to electric field is placed inbetween the vehicle chassis and the pair of power receiving plates. Thetime varying electric field sent by the sending plate 110 a, 120 a onthe roadway surface is received by the receiving plate 210 a, 210 b onthe vehicle chassis, to charge the onboard energy storage device 240.Because of the existence of the isolation material 220, the electricfield 130 is obstructed from being received by the vehicle chassis.

A cross-sectional view of the capacitive-coupled roadway power EV system100 is illustrated in FIGS. 16A and 16B. When an electric vehicle movesinto the electric field, the pair of power receiving plates 210 a, 210 bprovides a low impedance path for the electric field derived from theroadway sending plates 110 a, 110 b, and the power flow is transferredthrough the time varying electric field from the sending plate 110 a,110 b to the receiving plate 210 a, 210 b, which build up as twoparallel plate capacitors. The electric power received through theonboard plate 110 a, 110 b is delivered eventually to the onboard energystorage system 240 (e.g., battery packs). The electric power isgenerated from the power supply module 160 (e.g., an AC voltage isapplied to the pair of sending plates 210 a, 210 b). The voltagedifference applied on the pair of sending plates 210 a, 210 b iscontrolled as a constant value, therefore, for the scenario where two ormore EVs 130 are driving on one segment, each EV 130 can receive aninduced current on its receiving plate 110 a, 110 b to charge thevehicle battery 240. The charging status of one EV 130 will notinfluence the voltage difference applied on the pair of sending plates210 a, 210 b, which is controlled to be constant. In this way, thecharging status of one electric vehicle will not influence the powerreceiving status of another electric vehicle.

The operating frequency of capacitive-coupled roadway power EV system isthe dominant system parameter which affects the size, weight, cost andsystem efficiency. The system operating frequency should be much higherthan the universal industrial and household frequency (i.e., 50 or 60Hz). With higher operating frequency, the reactive power stored in thetwo parallel plate capacitors formed by the pair of sending andreceiving plates is larger with a fixed size of the plates, more energyis delivered to the EV. Conversely, to deliver sufficient electric powerto a moving EV 130, the system size is smaller by adapting higheroperating frequency. Although not limited hereto, the preferredoperating frequency of a capacitive-coupled roadway power EV system is20 kHz to 10 MHz. This operating frequency range is made withconsideration to both the size of the system and the ability of the highpower switching components.

With continued reference to FIG. 16A, it is seen that an insulationmaterial 310 a, 310 b with high electric resistivity is applied betweenthe road surface sending plate 110 a, 110 b and the roadway, for exampleconsisting mainly of asphalt 320. The roadway insulation material 310 a,310 b can prevent the electric field generating by the sending plate 110a, 110 b to be received by the metallic material in the asphalt 320. Ifthe electric vehicle 130 is not driving along the center line of theroadway, it can still receive a portion of the rated power capacity asseen in FIG. 16B.

In FIG. 17, there is shown a block diagram of the capacitive-coupledroadway power EV system 100 made in accordance with the presentdisclosure. The roadway power sending plates 110 a, 110 b receives theelectric power eventually from the power source 160, which is usually autility power distribution system, such as is provided by a publicutility company. Typically, the utility company provides electricalpower to most customers as 3-phase 60 Hz power, at 220 VAC. In thisembodiment, the power factor correction (PFC) unit 434 minimizes thereactive power generated by the system, and converts the 50 or 60 Hz ACvoltage to a DC voltage. The DC voltage is then converted to thedesirable voltage level by the voltage regulation unit 433. The DCvoltage is converted to high frequency AC voltage with the inverter 431.

The voltage across the pair of power sending plates 110 a, 110 b ismonitored and controlled to be constant. It is realized by the controlsystem 432 to detect the voltage difference on the two sending plate 110a, 110 b and then control the voltage regulation unit 433 to maintainthe constant voltage difference across the two sending plates 110 a, 110b.

The coupling between the pair of sending plates 110 a, 110 b and thepair of receiving plates 210 a, 210 b is referred to as capacitivecoupling by electric field. The sending and receiving plates act asparallel plate capacitors. It is not the same type of coupling thatoccurs in a transformer, which is an inductive coupling where theelectric power is stored and transferred using the magnetic field.Therefore, the magnetic field radiation to human tissue and otherelectronic devices for a capacitive-coupled roadway power EV system ismuch less than an inductive coupled system.

The electric power received on the two receiving plates 210 a, 210 b isAC current. It is turned to DC current by the rectifier unit 421 and theelectric filter 422. The DC current is then used to charge the onboardenergy storage device 240.

The onboard power meter 423 can accurately measure the received power ofthe electric vehicle. The power meter 423 is coupled to a display panelinside the EV, or can be assembled as part of the dashboard panel of theEV. It provides information to a driver of the vehicle such that thebattery information, charging status and cost of the received power, andthe like. The driver can turn on and off of the power receiving modulein the electric vehicle. In addition, the instantaneous power is alsodisplayed on the panel. If the instantaneous power is lower than therated value, this means that the vehicle is driving deviate from thecenter line of the roadway and the panel will send a warning message tothe driver when the power received is lower than the rated power, sothat the driver can adjust the position of the EV according to thewarning. In one embodiment, the instantaneous power information of thepower meter 423 is coupled to the automatic steering system that canadjust the EV's position with the assistant steering system.

Compensation network 430 comprising series connected, or parallel, orseries-parallel connected inductors and/or capacitors is electricallyconnected to one of the sending plates 120 a, to resonant with theequivalent capacitance between the sending plates 120 and the receivingplates 210, reducing the reactive power of the circuit. Compensationnetwork 420 comprising series connected, or parallel, or series-parallelconnected inductors and/or capacitors, electrically connected to one ofthe receiving plates 210 b is resonant with the equivalent capacitancebetween the sending plates 120 and the receiving plates 210, maximizingthe power transfer capability. In one embodiment, the compensationnetworks 420, 430 are implemented as either the LCLC topology or theCLLC topology described above. Other topologies are also contemplatedwith the broader aspects of the capacitive-coupled roadway powerelectric vehicle system.

The foregoing description of the embodiments has been provided forpurposes of illustration and description. It is not intended to beexhaustive or to limit the disclosure. Individual elements or featuresof a particular embodiment are generally not limited to that particularembodiment, but, where applicable, are interchangeable and can be usedin a selected embodiment, even if not specifically shown or described.The same may also be varied in many ways. Such variations are not to beregarded as a departure from the disclosure, and all such modificationsare intended to be included within the scope of the disclosure.

What is claimed is:
 1. A wireless power transfer system, comprising: apair of coupling capacitors, each coupling capacitor having an inputterminal and an output terminal; a send unit configured to transferpower capacitively through the pair of coupling capacitors, wherein thesend unit includes: an inverter configured to receive a DC input signaland convert the DC input signal to an AC input signal at a desiredresonant frequency; a send side compensation circuit interconnecting theinverter with the pair of coupling capacitors, wherein the send sidecompensation circuit is comprised of two bypass capacitors and eachbypass capacitor is connected in parallel between input terminals of thepair of coupling capacitors; and a receive unit configured to receivepower via the pair of coupling capacitors from the send unit, whereinthe receive unit includes a receive side converter configured to receivean AC charging signal from the pair of coupling capacitors and convertthe AC charging signal to a DC charging signal; and a receive sidecompensation circuit interconnecting the pair of coupling capacitorswith the receive side converter, wherein the receive side compensationcircuit is comprised of two bypass capacitors and each bypass capacitoris connected is connected in parallel between output terminals of thepair of coupling capacitors, wherein capacitance of each of the bypasscapacitors is at least five times larger than capacitance of each of thecoupling capacitors.
 2. The wireless power transfer system of claim 1wherein the four bypass capacitors are at least ten times larger thanthe coupling capacitors.
 3. The wireless power transfer system of claim1 wherein the send side compensation circuit is defined as two LCcircuits coupled in series.
 4. The wireless power transfer system ofclaim 3 wherein the receive side compensation circuit is symmetric withthe send side compensation circuit.
 5. The wireless power transfersystem of claim 1 wherein the inverter is further defined as a fullbridge converter circuit comprised of four switches or a half-bridgeconverter circuit comprised of two switches.
 6. The wireless powertransfer system of claim 5 further comprises a controller electricallycoupled to the four switches and operates to turn the switches on andoff at a zero voltage switching condition.
 7. The wireless powertransfer system of claim 1 wherein the receive side converter is furtherdefined as a full wave rectifier circuit.
 8. The wireless power transfersystem of claim 7 further comprises a battery configured to receive theDC charging signal from the receive side converter.
 9. The wirelesspower transfer system of claim 8 wherein the receive unit is integratedinto a vehicle.
 10. A compensation circuit for a capacitive powertransfer system having a first coupling capacitor and a second couplingcapacitor, comprising: a first bypass capacitor electrically coupled inparallel between input terminals of the first and second couplingcapacitors; a first inductor having an input terminal and an outputterminal, wherein the output terminal of the first inductor iselectrically coupled at a first node to the input terminal of the firstcoupling capacitor; a second inductor having an input terminal and anoutput terminal, wherein the output terminal of the second inductor iselectrically coupled at a second node to the input terminal of the firstinductor; and a second bypass capacitor electrically coupled in parallelwith the first bypass capacitor, wherein one terminal of the secondbypass capacitor is electrically coupled to the second node.
 11. Thecompensation circuit of claim 10 wherein capacitance of the first bypasscapacitor and capacitance of the second bypass capacitor are at leastfive times larger than capacitance of each of the coupling capacitors.12. The compensation circuit of claim 10 further comprises a thirdbypass capacitor electrically coupled in parallel between outputterminals of the first and second coupling capacitors; a third inductorhaving an input terminal and an output terminal, wherein the inputterminal of the third inductor is electrically coupled at a third nodeto the output terminal of the first coupling capacitor; a fourthinductor having an input terminal and an output terminal, wherein theinput terminal of the fourth inductor is electrically coupled at afourth node to the output terminal of the third inductor; and a fourthbypass capacitor electrically coupled in parallel with the third bypasscapacitor, wherein one terminal of the fourth bypass capacitor iselectrically coupled to the fourth node.
 13. A wireless power transfersystem, comprising: a pair of coupling capacitors, each couplingcapacitor having an input terminal and an output terminal; a send unitconfigured to transfer power capacitively through the pair of couplingcapacitors, wherein the send unit includes: an inverter configured toreceive a DC input signal and operates to convert the DC input signal toan AC input signal at a desired resonant frequency; a send sidecompensation circuit interconnecting the inverter with the pair ofcoupling capacitors, wherein the send side compensation circuitincludes: a first bypass capacitor electrically coupled in parallelbetween input terminals of the pair of coupling capacitors; a firstinductor having an input terminal and an output terminal, wherein theoutput terminal of the first inductor is electrically coupled at a firstnode to the input terminal of one of the pair of coupling capacitors; asecond inductor having an input terminal and an output terminal, whereinthe output terminal of the second inductor is electrically coupled at asecond node to the input terminal of the first inductor; and a secondbypass capacitor electrically coupled in parallel with the first bypasscapacitor, wherein one terminal of the second bypass capacitor iselectrically coupled to the second node.
 14. The wireless power transfersystem of claim 13 wherein capacitance of the first bypass capacitor andcapacitance of the second bypass capacitor are at least five timeslarger than capacitance of each of the coupling capacitors.
 15. Thewireless power transfer system of claim 13 wherein the inverter isfurther defined as a full bridge converter circuit comprised of fourswitches or a half-bridge converter circuit comprised of two switches.16. The wireless power transfer system of claim 15 further comprises acontroller electrically coupled to the four switches and operates toturn the switches on and off at a zero voltage switching condition. 17.The wireless power transfer system of claim 13 further comprises areceive unit configured to receive power via the pair of couplingcapacitors from the send unit, wherein the receive unit includes areceive side converter configured to receive an AC charging signal fromthe pair of coupling capacitors and convert the AC charging signal to aDC charging signal; and a receive side compensation circuitinterconnecting the receive coil with the receive side converter. 18.The wireless power transfer system of claim 17 wherein the receive sideconverter is further defined as a full wave rectifier circuit.
 19. Thewireless power transfer system of claim 17 wherein the receive unit isintegrated into a vehicle.
 20. The wireless power transfer system ofclaim 17 wherein the receive side compensation circuit includes a thirdbypass capacitor electrically coupled in parallel between outputterminals of the first and second coupling capacitors; a third inductorhaving an input terminal and an output terminal, wherein the inputterminal of the third inductor is electrically coupled at a third nodeto the output terminal of the first coupling capacitor; a fourthinductor having an input terminal and an output terminal, wherein theinput terminal of the fourth inductor is electrically coupled at afourth node to the output terminal of the third inductor; and a fourthbypass capacitor electrically coupled in parallel with the third bypasscapacitor, wherein one terminal of the fourth bypass capacitor iselectrically coupled to the fourth node.